Apparatus and method for clock synchronization in a multi-point OFDM/DMT digital communications system

ABSTRACT

A multi-point communications system is set forth herein. The communications system comprises a transmitter for transmitting OFDM/DMT symbols over a predetermined number of bins across a transmission medium. The OFDM/DMT symbols are generated using at least one timing signal. At least one of the predetermined number of bins includes a pilot tone sub-symbol having a frequency corresponding to the clock signal. The communications system also includes a receiver for receiving the OFDM/DMT symbols from the transmission medium. The receiver demodulates the received symbols using at least one timing signal. The receiver has a first pilot tone search mode of operation in which the receiver adjusts its timing signal to scan the frequency range of the predetermined number of bins looking for the pilot tone sub-symbol and identifies the bin including the pilot tone sub-symbol. The receiver further has a subsequent second pilot tone acquisition mode in which the receiver adjusts the timing signal to receive the identified bin containing the pilot tone sub-symbol and measures phase differences between successive pilot tone sub-symbols to thereby perform a further adjustment of the timing signal so that the pilot tone sub-symbol is received within a frequency range sufficient for subsequent phase locked loop processing thereof.

The present application is a continuation of U.S. application Ser. No.10/813,695, filed on Mar. 29, 2004, which is a continuation of U.S.application Ser. No. 09/665,225, filed on Sep. 18, 2000, now U.S. Pat.No. 6,771,590 which is a continuation of U.S. application Ser. No.08/900,312, filed on Jul. 25, 1997, now U.S Pat. No. 6,122,246 which isa continuation-in-part of U.S. application Ser. No. 08/700,779, filedAug. 22, 1996, now U.S. Pat No. 5,790,514 and a continuation-in-part ofU.S. application Ser. No. 08/845,544, filed Apr. 24, 1997 now U.S. Pat.No. 6,285,654.

The entire teachings of the above applications are incorporated hereinby reference.

BACKGROUND OF THE INVENTION

The present invention is directed to an OFDM/DMT digital communicationssystem. More particularly, the present invention is directed to anapparatus and method for synchronizing the clocks used in a transmitterand receiver of an OFDM/DMT digital communications system. The presentinvention is particularly applicable in multipoint OFDM/DMT digitalcommunications systems.

Multipoint communications systems having a primary site that is coupledfor communication with a plurality of secondary sites are known. Onesuch communications system type is a cable telephony system. Cabletelephony systems transmit and receive telephone call communicationsover the same cable transmission media as used to receive cabletelevision signals and other cable services.

One cable telephony system currently deployed and in commercial use isthe Cablespan 2300 system available from Tellabs, Inc. The Cablespan2300 system uses a head end unit that includes a primary transmitter andprimary receiver disposed at a primary site. The head end unit transmitsand receives telephony data to and from a plurality of remote serviceunits that are located at respective secondary sites. This communicationscheme uses TDM QPSK modulation for the data communications and canaccommodate approximately thirty phone calls within the 1.9 MHzbandwidth typically allocated for such communications.

As the number of cable telephony subscribers increases over time, theincreased use will strain the limited bandwidth allocated to the cabletelephony system. Generally stated, there are two potential solutions tothis bandwidth allocation problem that may be used separately or inconjunction with one another. First, the bandwidth allocated to cabletelephony communications may be increased. Second, the availablebandwidth may be used more efficiently. It is often impractical toincrease the bandwidth allocated to the cable telephony system given thecompetition between services for the total bandwidth available to thecable service provider. Therefore, it is preferable to use the allocatedbandwidth in a more efficient manner. One way in which the assignedbandwidth may be used more efficiently is to use a modulation schemethat is capable of transmitting more information within a givenbandwidth than the TDM QPSK modulation scheme presently employed.

The present inventors have recognized that OFDM/DMT modulation schemesmay provide such an increase in transmitted information for a givenbandwidth. Such systems, however, present a number of technicalproblems. One such problem is the determination of how one or moreremote receivers are to synchronize their internal clocks and timingsystems with the internal clock and timing system of a primarytransmitter at a central site. A remote receiver must first synchronizeits internal clock and timing system with the clock used by the primarytransmitter to synthesize the transmitted signal before the remotereceiver can properly demodulate the data that it receives. A furtherproblem occurs in multipoint communication systems in which there areplural groups of remote transmitters that transmit data to centralizedtransceivers. Each group of transmitters often has its transmissionsfrequency multiplexed with transmissions from other groups before beingdemultiplexed for receipt by a particular central transceiver. Theresulting multiplexing/demultiplexing operations introduce frequencyoffsets for which compensation must be made if the receiver of thecentral transceiver is to properly extract the correct data from thesignals that is receives. The present inventors have recognized the needfor such upstream and downstream clock synchronization and havedisclosed solutions to these problems.

BRIEF SUMMARY OF THE INVENTION

A multi-point communications system is set forth herein. Thecommunications system comprises a transmitter for transmitting OFDM/DMTsymbols over a predetermined number of bins across a transmissionmedium. The OFDM/DMT symbols are generated using at least one timingsignal. At least one of the predetermined number of bins includes pilottone sub-symbols generated from a pilot tone having a frequencycorresponding to the at least one timing signal. The communicationssystem also includes a receiver for receiving the OFDM/DMT symbols fromthe transmission medium. The receiver demodulates the received symbolsusing at least one timing signal. The receiver has a first pilot tonesearch mode of operation in which the receiver adjusts its timing signalto scan the frequency range of the predetermined number of bins lookingfor the pilot tone sub-symbols and identifies the bin including thepilot tone sub-symbols. The receiver further has a subsequent secondpilot tone acquisition mode in which the receiver adjusts the timingsignal to receive the identified bin containing the pilot tonesub-symbol and measures phase differences between successive pilot tonesub-symbols to thereby perform a further adjustment of the timing signalso that the pilot tone sub-symbol is received within a frequency rangesufficient for subsequent phase locked loop processing thereof.

In accordance with one advantageous embodiment of the system, the timingsignal of the transmitter is used for timing inverse Fourier transformprocessing and for carrier generation in transmitting the OFDM/DMTsymbols while the timing signal of the receiver is used for timingFourier transform processing and for carrier generation in demodulatingthe received OFDM/DMT symbols.

Other features and advantages of the present invention will becomeapparent upon review of the following detailed description andaccompanying drawings.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

FIG. 1 is a schematic block diagram of a multi-point communicationssystem having a plurality of remote service units disposed at aplurality of secondary sites wherein each of the remote service unitscomprises a receiver having an improved receiver architecture.

FIG. 2 illustrates two symbol constellations that are transmitted in twoseparate frequency bins in accordance with OFDM/DMT data modulationtechniques.

FIG. 3 is a block diagram of one embodiment of a head end unit and aremote service unit of the communications system of FIG. 1 showing thosecomponents involved in downstream synchronization.

FIG. 4 illustrates various spectral distributions for the pilot tone binand adjacent bins.

FIG. 5 is a flow chart illustrating one manner of executing the firstpilot tone search mode of receiver operation.

FIG. 6 is a flow chart illustrating one manner of executing the secondpilot tone acquisition mode of receiver operation.

FIG. 7 is a block diagram of on embodiment of a head end unit and aremote service unit of the communications system of FIG. 1 showing thosecomponents involved in upstream synchronization.

FIG. 8 is a flow chart illustrating one manner of executing the upstreamsynchronization.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 is a block diagram of a multi-point communications system whichmay use a remote service unit having the improved receiver andtransmitter architectures disclosed herein. As illustrated, thecommunications system, shown generally at 20 includes a head end unit(HE) 25 disposed at a primary site. The head end unit communicates witha plurality of remote service units (RSUs) 30 respectively disposed at aplurality of secondary sites, over a transmission medium 35 such as acoaxial cable.

The digital communications system 20 may, for example, be a cabletelephony system. In such an application, the head end unit 25 isdisposed at a cable television transmission facility while the remoteservice units 30 are disposed at individual customer locations, such asindividual customer homes. The transmission medium 35 would be the newor existing transmission cable used to transmit the cable televisionservices. The head end unit 25 in a cable telephony network isresponsible for communicating with and interconnecting telephone callsbetween the plurality of remote service units 30 as well ascommunicating with a central switching office 40 for sending andreceiving telephone calls from sites exterior to the local cabletelevision service area.

The present system 20 utilizes OFDM/DMT digital data modulation forexchanging communications data between the head end unit 25 and theremote service units 30. Such OFDM/DMT digital data communicationsassign a particular amplitude, frequency, and phase for each transmitted“sub-symbol”. The transmitted “sub-symbol” represents one or moreinformation data bits that are to be transmitted between the units 25and 30. Each sub-symbol may be represented by a point within a“constellation”, the point being transmitted at a given carrierfrequency or “bin”.

FIG. 2 illustrates the use of two constellations 90 and 95, each havingsixteen constellation points that are capable of being transmittedwithin two separate frequency bins. As illustrated, a sub-symbol havinga carrier signal of frequency f₁ has its amplitude and phase varieddepending on the constellation point that is to be transmitted. Forexample, a constellation point representing the binary states 0000 istransmitted as a sub-symbol at a phase of θ₁ and an amplitude of A₁during a designated symbol time. A constellation point representing thebinary states 1111, however, is transmitted as a sub-symbol at a phaseof θ₂ and an amplitude of A₂ during a designated symbol time. Similarly,the second constellation 95, preferably having the same amplitude andphase designations for its sub-symbols as the first constellation 90, isused to modulate a second carrier frequency f₂. The resulting modulatedsignals are combined into a single output symbol in which the individualsub-symbols are differentiated from one another based on theirrespective carrier frequencies or “bins”. It will be recognized thatmany variations of the disclosed OFDM/DMT transmission scheme arepossible, the foregoing scheme being merely illustrated herein toprovide a basic understanding of OFDM/DMT communications.

A block diagram of one embodiment of the general components of atransmitter 97 of a head end unit 25 and a receiver 150 of a remoteservice unit 30 used for downstream clock synchronization is shown inFIG. 3. As illustrated, the transmitter 97 of the head end unit 25receives data at one or more lines 105 and supplies this data to adigital signal processor 110 and its associated components(collectively, DSP). The DSP 110 accepts the digital data and performs aFourier Transform, preferably an Inverse Fast Fourier Transform (IFFT),on the received data. The digital data resulting from the IFFTtransformation is provided to the input of a digital-to-analog converter115. The analog signal resulting from the conversion is preferablyprovided to the input of a band pass filter 120, the output of which issupplied to the input of a mixer 125 where the filtered output signal atline 130 is mixed to a first frequency f_(x). A further band pass filter135 is provided to remove the images resulting from the mixing process.The filtered output signal at line 140 is subject to a further mixing atmixer 145 where it is mixed to a second frequency f_(y) suitable fortransmission along the transmission medium 35 (e.g., RF, opticalfrequencies, etc.).

The receivers 150 of the remote service units 30 receive the OFDM/DMTdata from the transmission medium 35. The received signal is demodulatedusing a first frequency, preferably f_(y), at mixer 160. The mixedsignal is provided to the input of a first bandpass filter 165 thatfilters the images resulting from the mixing process. The resultingsignal is further mixed with a demodulating signal f_(x) to a basebandlevel for the receiver 150 at mixer 170. Again, the images resultingfrom the mixing process are removed by a bandpass filter 175. Thefiltered signal is applied to the input of an analog-to-digitalconverter 180 that converts the filtered analog signal to digitalsamples that are subsequently processed by the digital signal processingportions 185 of the receiver 150. In that processing, the digital datasignals received from the analog-to-digital converter 180 undergo aFourier Transform, preferably an FFT, to extract the frequency and phasecomponents of the received signal. Based on this processing, the digitalsignal processing portions 185 may reconstruct the data provided to thetransmitter 97 at lines 105 of the head end unit 25.

As illustrated in FIG. 3, the transmitter 97 of the head end unit 25includes a voltage controlled oscillator 190. The voltage controlledoscillator 190 provides a common clocking signal at lines 200 inresponse to a signal received from the DSP 110 or an analog phase lockedloop signal 192 received on a back plane. The clocking signal at lines200 is provided to, for example, the input of a digital counter 205 thatgenerates a further clocking signal at line 210 that is provided to theinput of the digital-to-analog converter 115 to control timing of theconversion process. This same clocking signal output 200 from thevoltage controlled oscillator 190 is provided to the input of frequencysynthesizer 215 and frequency synthesizer 220. The frequencysynthesizers 215 and 220 use the received clocking signal to generatethe mixing signals f_(x) and f_(y) to mixers 125 and 145, respectively.Thus, the timing of the OFDM/DMT symbol generation and the mixing of thegenerated symbols for transmission along transmission medium 35 aredependent upon the frequency and phase of the output signal 200 of thevoltage controlled oscillator 190.

As noted above in the description of the receiver 150, the receiver 150executes several mixing operations on the received signal and, further,performs an analog-to-digital conversion of the received signal. Toproperly perform the mixing and conversion operations and ensure theintegrity of the extracted OFDM/DMT data, it is desirable to synchronizethe signals used for the generation of the OFDM/DMT transmission by thetransmitter 97 with the signals used for the demodulation and conversionof the symbols received at the receiver 150. Applying this principal tothe embodiment set forth in FIG. 3, the signals output by frequencysynthesizers 215 and 220 to mixers 125 and 145 at the transmitter 97should be synchronized with the signals output by the frequencysynthesizers 230 and 235 to mixers 160 and 170 at the receiver 150.Similarly, the clocking signal 210 supplied to the digital-to-analogconverter 115 should be synchronized to the sampling clock signal 237supplied to the analog-to-digital converter 180.

In the disclosed embodiment, a reference clock signal is embedded in thebaseband OFDM digital signal by the transmitter 97 in order to performthe desired synchronization. This reference clock signal takes the formof a constant amplitude and phase sub-symbol that is transmitted in aparticular frequency bin, and is called the pilot tone sub-symbol. Thepilot tone sub-symbol has a frequency and phase corresponding to thesignal output 200 from the voltage controlled oscillator 190 of thetransmitter 97. Preferably, several bins are designated to include thepilot tone sub-symbol. Each receiver 150 seeks to recover any one ofthese tones, with the remaining pilot tones designated as backups. Thereceivers 150 demodulate the RF passband signal assigned to the RSU andattempt to extract the pilot tone from the baseband signal. The eventualgoal is to “lock on” to the exact phase and frequency of the pilot tone.This is done by appropriately adjusting a voltage controlled oscillator240 of the receiver 150 using a phase locked loop, such as a digitalphase locked loop (DPLL). This results in the locking of the samplingclock signal 237 (A/D sampling rate) with the clocking signal 210 usedby the transmitter 97 to control the timing of the D/A conversion of thebinary data received at lines 105. Since the RF carrier frequency andsample clocks are tied together, this circuit topology and correspondingmethod also simultaneously accomplishes carrier recovery.

The entire pilot tone acquisition procedure can be viewed as a two stageprocess comprising search and acquisition of the pilot tone. To thisend, the receiver 150 operates in accordance with at least two modes ofoperation. In a first pilot tone search mode of operation, the receiverscans the frequency range of the bins transmitted by the transmitter 97in predetermined frequency steps looking for the bin containing thepilot tone. Once the bin containing the pilot tone sub-symbol has beenidentified, the receiver 150 makes a gross timing adjustment of theoutput signal of voltage controlled oscillator 240 to receive the binincluding the pilot tone sub-symbol in the correct predetermined binlocation. In a subsequently occurring second pilot tone acquisitionmode, the receiver 150 also measures the phase difference betweenconsecutive pilot tone sub-symbols to adjust the timing of the output ofthe voltage controlled oscillator 240 so that it is within a frequencyrange sufficient for subsequent phase locked loop processing of thepilot tone signal. After this acquisition has taken place, the receiver150 switches to a steady state tracking mode in which the phase lockedloop is used to constantly maintain synchronism with the transmitter 97.

There are at least two ways in which this subsequent synchronism may bemaintained. First, when the PLL is using the pilot-tone, the constantpilot-tone sub-symbol is known to the PLL beforehand. For each symboltime, the receiver demodulates the pilot-tone bin and computes the errorbetween the demodulated sub-symbol and the known value. This errorsignal then is filtered and run through the D/A to control the VCXOappropriately. In contrast, a decision directed mode may also the usedto maintain synchronism. When decision directed mode is used, the PLLselects a some bin carrying random data for processing. The receiverdoes not know the transmitted sub-symbols a priori. Each symbol time,therefore, the chosen bin is demodulated and then sliced to the nearestconstellation point. The difference (or error) between the sliced andun-sliced sub-symbol is used to drive the PLL as before. Operation inthe decision directed mode is limited to situations in which thedecisions are expected to be correct.

The foregoing, two-stage operation of the receiver 150 is used at “coldstart-up” of the receiver 150. This happens, for instance, when an RSU30 powers up for the first time or when it attempts to re-establishcommunication with the HE 25 after a prolonged period of inactivity. Theacquisition process follows a successful search. However, the firstpilot tone search mode in which the receiver 150 performs the requisitesearch for the bin containing the pilot tone sub-symbol may be skippedfor “warm start-ups”, i.e. when the receiver 150 already has a fix onthe location of the pilot tone subsymbol and is merely attempting tore-establish communication with the HE 25 after a brief period ofdisruption. After successful acquisition of the pilot tone in the secondpilot tone acquisition mode, a steady state tracking procedure isinitiated by the receiver 150 that thereafter maintains the timing ofthe RSU 30 in synchronism with the timing of the HE 25.

The foregoing modes of receiver operation are desirable for a number ofreasons. One reason relates to the result of discrepancies between theoutput signals of the voltage controlled oscillators 190 and 240 at theHE transmitter 97 and RSU receiver 150, respectively. Such signalspotentially have frequency deviations (typically specified in ppm) that,when coupled with the high RF modulating frequencies typically used fortransmission over the transmission medium 35, result in mutual frequencyoffsets. For example, if the voltage controlled oscillators 190 and 240are specified at 30 ppm and 50 ppm respectively, then for an RFmodulating frequency of 750 MHz, frequency offsets up to +/−60 KHz arepossible. As a result of this frequency offset, the downconverted OFDMbaseband spectrum at the receiver 150 is displaced in frequency. Thefrequency locking range of a typical baseband DPLL is in the order of500 Hz and, thus, the pilot tone will most likely lie outside thefrequency locking range of the DPLL. The foregoing receiver modeoperations bring the pilot tone within the locking range of such a DPLL.

As noted above, two receiver modes of operation are employed. The firstpilot tone search mode of operation performs a generally grossadjustment of the voltage controlled oscillator 240 that reduces thefrequency offset between the pilot tone and the output of the voltagecontrolled oscillator 240 to within a fraction of a frequency bin (e.g.,½ to ¼ of a bin). The second pilot tone acquisition mode of operationuses the gross adjustment to provide a more accurate estimate of thefrequency offset which is then used as the starting point for the DPLL.

To begin the first pilot tone search mode of operation, the RSU receiver150 A/D samples the baseband OFDM signal at the analog-to-digitalconverter 180 and inputs the samples into, for example, a hardwarecorrelator 250. Details of one embodiment of such a hardware correlator,although not pertinent to the present invention, can be found in U.S.Ser. No. 08/845,544, filed Apr. 24, 1997. The correlator 250 can beviewed as a direct digital implementation of a discrete Fouriertransform (DFT) over, for example, 9 frequency bins. In other words, thecorrelator 250 provides symbol rate complex outputs to the DSP 255 forsubsequent processing. In the present embodiment, a substantial portionof the first and second receiver modes of operation are implemented inthe DSP 255 on symbol rate outputs from the hardware correlator 250.

In accordance with one embodiment of the first pilot tone search mode ofreceiver operation, the number of bins that need to be searched has tobe determined. Generally, such a determination depends upon theparticular system requirements. For exemplary purposes, it will beassumed that the system employs a bin width of 9.615 KHz and uses apilot tone bin that is located as the 12^(th) bin in a total receivewindow of 26 bins. The number of bins to be searched depends on theaccuracy of the voltage controlled oscillators 190 and 240 at the HE 25and RSU 30. If frequency offsets of up to +/−60 KHz are possible (seeabove), and each bin is 9.615 KHz wide, 13 bins will be searched—the12^(th) bin and 6 bins on each side of the 12^(th) bin in the window.

It is also desirable to select a predetermined spectral pattern thatwill be used to find the pilot tone. As such, a sufficient and minimalset of metrics based on the pattern that will indicate with highprobability that the pilot is found may be employed in the first pilottone search mode. This pattern should be limited to as few bins aspossible to minimize bandwidth usage while at the same time beingdistinct from the rest of the spectrum. Several possible symmetric andasymmetric spectral patterns are shown in FIGS. 4( a)-4(d).

For exemplary purposes, the 5-bin-wide pattern of FIG. 4( a) is employedin the instant case. This pattern has a pilot tone disposed in one binwith two adjacent empty (zero power) bins on each side thereof. The twooutermost bins which are beside each of the empty bins carry eitheractual data or random data. This ensures that there will always be astable non-zero average power in these two outermost bins. Note that thewider the pattern is, the larger the number of bins that need to bereceived. For instance, if 13 bins are possible candidates for the pilotlocation, then this 5-bin pattern requires that the receiver 150 receiveand process 17 bins.

It is further assumed for purposes of the present example that there areeight, 26-bin windows transmitted by the HE 25, and that fourpredetermined windows among the eight will include predetermined binshaving the spectral pattern with the pilot tone. Pursuant to the firstpilot tone search mode of operation, the receiver 150 of the RSU 30adjusts to receive one of the predetermined windows with the pilot tone.A selected window number is predetermined and back-up windows are alsopredetermined in the event that pilot recovery fails for the window ofchoice.

FIG. 5 illustrates one manner of implementing the first pilot tonesearch mode in the receiver 150. In the specific implementationillustrated here, a wide range of tests are performed on receivedsignals to verify the presence of the pilot tone. However, it will berecognized that fewer than all such tests may be used in this mode yetstill achieve satisfactory results.

In connection with the search mode, each RSU 30 has a table of, forexample, five frequency offset values (also called DAC offset table) forits voltage controlled oscillator 240 for each window. The DAC table isused to scan a window in, for example, five steps of 1.923 KHz each(9615/5 Hz). By using scanning steps equal to a fifth of the width of abin, it is possible get an estimate of the pilot location to within1.923 KHz (9615/5 Hz). To implement the scanning, the receiver 150 ofthe RSU 30 runs through a table of five output values supplied to thedigital-to-analog converter 265. The output of the digital-to-analogconverter 265 is supplied to the input of a low pass filter 270, theoutput of which is an analog control voltage that alters the frequencyof the output signal from the voltage controlled oscillator 240. Thus,each value that is output corresponds to a frequency offset of thevoltage controlled oscillator 240. For each offset value, the RSU 30carries out a number of functions.

First, the receiver 150 changes the frequency offset of the voltagecontrolled oscillator 240 by a fifth of the width of a bin. This changeof the frequency offset is performed by writing the correct DAC outputvalue in the table of five DAC output values to the digital-to-analogconverter 265. Immediately after writing to the digital-to-analogconverter 265, several symbols have to be discarded to allow thefrequency output of the voltage controlled oscillator 240 to stabilize.The number of symbols to be discarded will depend on the voltagecontrolled oscillator properties and the magnitude of the change infrequency. As such, the receiver 150 may need to discard a few hundredsymbols.

Next, a complex phase correction term due to the frequency offset in thereceiver 150 is applied to each received sub-symbol. Such a phasecorrection is necessary when the data transmitted by the transmitter 97is in the form of a formatted data frame having, for example, a cyclicprefix or an analogous counterpart. The phase is corrected in accordancewith the following equation:J _(i,j)(l)=I _(i,j)(l)e ^(√{square root over (−1)}Φl)For i=1, . . . , 17 and j=0, 1, 2, 3, 4 and where

$\Phi = {2\;\pi\;{f_{off}( \frac{CP}{N} )}}$is the incremental phase correction applied to all received sub-symbolsin the chosen window received with an offset of f_(off) frequency binswith respect to the transmitted bins (see the discussion in thefollowing paragraph), and l is the symbol index. Here, the j indexcorresponds to the index into the DAC offset table corresponding to thefrequency steps used to scan the predetermined window having the pilottone. The term I_(i,j)(l) is the received symbol in the i^(th) bin forthe j^(th) DAC offset table value.

This incremental phase shift is applied in systems in which a cyclicprefix is used and each of the remote receivers are designed to onlyprocess signals within a passband beginning at a fixed frequencycorresponded to a predetermined bin (e.g., bin 64). In such a system,the HE transmits across substantially the entire available signalspectrum. However, the receiver of each RSU mixes the transmitted signalso that the frequency range of the window that it is to process is mixedto begin at the fixed frequency corresponding to the predetermined bin.The f_(off) values for each window, accordingly, are system dependentand, thus, are dependent on the system design parameters. It should benoted that the value of CP is zero if a cyclic prefix, or an analogouscounterpart, is not transmitted from the transmitter 97.

After the phase correction has been completed, the power of the signalreceived in each of the bins is computed as is the signal's correlation.Assuming that the pilot signal bin is known to be the 12th bin in thewindow, the squared magnitude of each symbol in the 12th bin in thewindow and the 8 bins on either side of the 12th bin (for a total of 17bins) is computed.

The receiver 150 computes the correlation of the current symbolJ_(i,j)(l) with the complex conjugate of the previous symbolJ_(i,j)(l−1) for the 12th bin and 6 bins on each side of the 12th bin(for a total of 13 bins) in accordance with the following equation:R _(i,j)(l)=J _(i,j)(l)×J _(i,j)*(l−1)where i is the bin number and j is the index into the DAC offset tablefor the voltage controlled oscillator 240 for each of these equations.The correlation will be constant if the bin number and index includesthe pilot tone.

The foregoing power and correlation calculations are then averaged foreach bin i and index j over L symbols. Accordingly, the foregoingoperations are repeated for L symbols to compute the averages inaccordance with the following averaging equations:

${P_{i,j}(L)} = {\frac{1}{L}{\sum\limits_{l = 1}^{L}\;{{J_{i,j}(l)}}^{2}}}$${H_{i,j}(L)} = {{\frac{1}{L}{\sum\limits_{l = 1}^{L}\;{R_{i,j}(l)}}}}^{2}$where P_(i,j) and H_(i,j) are the average power and coherence in the ithbin and for the jth index over L symbols.

After averaging, the metrics for each bin i and index j are computed. Tothis end, for each index j, the following metrics are computed for the12th bin and 6 bins on each side of the 12th bin (for a total of 13bins):A _(i,j) =P _(i,j)(L)−P _(i−1,j)(L)−P _(i+1,j)(L)B _(i,j) =|P _(i−1,j)(L)−P _(i+1,j)(L)|C _(1,i,j) =P _(i−2,j)(L)−P _(i−1,j)(L)C _(2,i,j) =P _(i+2,j)(L)−P _(i+1,j)(L)

Note that the number of metrics and the definition of the metrics willdepend on the spectral patterns used, the foregoing metrics being thoseused for the spectral pattern of FIG. 4( a).

If the table of output values to the voltage controlled oscillator 240is not exhausted after computing the metrics, the entire process aboveis repeated using the next value in the DAC offset table. If the tablehas been exhausted, H_(i,j)(L) and P_(i,j)(L) are re-ordered in apredetermined fashion. Where the DAC offset index runs from 0 to 4,H_(i,j)(L) is re-ordered as H_(i,0)(L), H_(i,1)(L), H_(i,2)(L), . . . .The same re-ordering is performed on array P_(i,j)(L). Such ordering canbe incorporated into earlier operation to save processing time by usingappropriate addressing methods. After this rearrangement, each of thearrays should have 65 elements.

The square root of each element H_(i,j)(L) in the array of correlationmetrics is then computed and the result is divided by the correspondingaverage power value P_(i,j)(L) to provide a normalized correlationmetric T_(i,j) in accordance with the following equation:

$T_{i,j} = \frac{\sqrt{H_{i,j}(L)}}{P_{i,j}(L)}$

The pilot tone will have a high degree of coherence and hence theT_(i,j) values will be large (close to 1) for bins in the neighborhoodof the bin containing the pilot tone. All other bins will have very lowcoherence (close to 0) as they will either be carrying uncorrelated dataor will have zero power and be effectively Gaussian noise. Thus, thecoherence test is a simple threshold test which eliminates fromconsideration all bins that have T_(i,j) values less than apredetermined value. This predetermined value, for example, may be inthe range of 0.5-0.8. The test can thus be represented as follows:Reject (bin, index) pair (i,j) if T_(i,j<λ where) 0.5<λ<0.8.

If all the bins in the window are eliminated, then the receiver 150 willmove on to a backup window and repeat the foregoing search process.

In addition to undergoing a coherence test, the acquired signals mayalso undergo an excessive coherence test that is executed during thefirst pilot tone search mode. This test is used to reject a window ifthere are coherent interferers too close to the pilot tone and/or toomany coherent interferers in the window. This test may be used tosuccessfully reject video channels when hands-off provisioning isdesired. In NTSB standard video channels, the spectrum has severalcarriers that are spaced evenly at about 16 KHz apart. This is veryclose to the bin spacing that is used in the exemplary embodiment here.If only the foregoing coherence test and the pattern matching tests(described below) were used, several bin groupings would pass thesetests. Thus, the test for excessive coherence is useful to ensure thatthere is only one grouping of bins that passes the tests.

The excessive coherence test involves using a lowpass filter with, forexample, 13 taps with all taps set to unity to filter the array ofT_(i,j) values. The maximum output of the filter, T_(max), is comparedto a threshold value T_(th) that is determined, for example,experimentally. If T_(max)<T_(th), then the window is rejected and thereceiver will move on to the next window and begin processing anew. Notethat the evenly spaced spectral pattern of the cable channel could berejected with either of the asymmetric spectral patterns in FIGS. 4(c)-4(d) with appropriate pattern matching tests. However, the excessivecoherence test is still useful in the case of coherent interference nearthe pilot tone.

The receiver 150 may then perform one or more successive patternmatching tests. The pattern matching tests compare the spectral patternaround each bin with the expected pattern in FIG. 4( a). The particularpattern matching tests depend on which pattern is used. In accordancewith a first test, the receiver rejects (bin,index) pairs that satisfythe following criterion:C_(1,i,j)<0 or C_(2,i,j)<0

Another possible rejection criterion that can be used instead of theabove isC_(1,i,j)<B_(i,j) or C_(2,i,j)<B_(i,j)

Again, if all the bins in the window are rejected, then the receiverwill move on to a backup window and start anew.

A further subsequent pattern matching test may also be performed by thereceiver 150. In accordance with this further test, the receiver rejectsthe bin/index pairs that satisfy the following criterion:A_(i,j)<μB_(i,j)

The value of μ is, for example, selected experimentally and usually liesbetween 1 and 10. If all the bins in the window are eliminated, then thereceiver will move on to a backup window and start anew.

After the above tests are performed, all the bins except for a fewgrouped around the bin containing the pilot tone will be rejected. Thebest bin i and index j into the DAC offset table for the voltagecontrolled oscillator 240 is found by searching for the largest T_(i,j)value among the 65 values. Using the index j corresponding to thelargest T_(i,j), the output value that is to be provided to thedigital-to-analog converter 265 is found from the table of DAC outputvalues and the voltage-controlled-oscillator is directed to proceed tothe correct position to allow further processing in the second pilottone acquisition mode of operation.

One embodiment of the implementation of the second pilot toneacquisition mode is set forth in connection with FIG. 6. With respect tothe second pilot tone acquisition mode, it can be shown that in theabsence of any interference from data carrying bins, there is a phaseoffset (in radians) between

$\Theta = {{2\;{\pi( {ɛ + {( {f_{off} + ɛ} )( \frac{CP}{N} )}} )}} = {{2\;\pi\;{ɛ( {1 + \frac{CP}{N}} )}} + \Phi}}$consecutive pilot symbols given by the equation:where ε is the fractional (normalized) frequency offset in bins. Forexample, if the residual frequency offset after the first pilot tonesearch mode is completed is 1 KHz, and for a bin width of 9.6 KHz,ε=1/9.6=0.104. The phase rotation Φ is predetermined by the windownumber and is corrected as soon as the correlator outputs are availableto the DSP. In the second pilot tone acquisition mode, the receiver 150extracts the phase differences between consecutive pilot tonesub-symbols to estimate ε. To this end, let I(n) denote the nth complexsymbol obtained after the phase correction noted above. To estimate thefrequency offset, the receiver 150 forms a sum using N symbols as,

${S(N)} = {\sum\limits_{l = 1}^{N}\;{{J(l)}{J^{*}( {l - 1} )}}}$where J*(n) denotes the complex conjugate of the nth symbol. An estimateof the fractional frequency offset is then obtained as

$\hat{ɛ} = {\frac{1}{2\;{\pi( {1 + \frac{CP}{N}} )}}a\;{\tan( \frac{{Im}( {S(N)} )}{{Re}( {S(N)} )} )}}$

Since {circumflex over (ε)} is a signed dimensionless number normalizedby the bin width, it is converted to a frequency offset (in Hertz) bymultiplying it by the bin width. This frequency offset is translated tothe appropriate numerical value and written out to the digital-to-analogconverter 265. To meaningfully interpret the output of the second pilottone acquisition mode, |{circumflex over (ε)}|<0.5, i.e. the magnitudeof the residual frequency offset is less than half of a bin width.

After the receiver 150 has completed operation in the second pilot toneacquisition mode, the voltage controlled oscillator 240 is set to thebest estimate of the pilot tone and is within the locking bandwidth of aDPLL. The DPLL begins operation by utilizing the difference between thephase of the received pilot tone and its constant desired (transmitted)value as the error signal. This symbol rate error signal is filtered by,for example, a type II proportional-integral (PI) loop filter togenerate the control voltage for adjusting the voltage controlledoscillator 240. The second order loop filter ensures a zero steady stateerror in the face of frequency offsets. The loop filter transferfunction is given by,

${L(z)} = {K_{P} + \frac{{zK}_{I}}{z - 1}}$which results in an overall transfer function of the form.

${H(z)} = \frac{{( {K_{P} + K_{I}} )z} - K_{P}}{z^{2} + {( {K_{P} + K_{I} - 2} )z} + 1 - K_{P}}$

With appropriately chosen DPLL parameters, phase lock can be achievedwithin a few thousand symbols. Also, several sets of DPLL parameters arechosen to provide multiple loop bandwidths. For instance, the loopbandwidth takes on the highest value during acquisition and the lowestduring tracking.

After sample timing has been acquired, the RSU receiver 150 constantlymaintains the correct timing. This requires the continuous operation ofthe DPLL. Since each RSU 30 in the present embodiment can demodulate atmost 9 bins, it is wasteful of bandwidth to constantly use these binsfor a pilot tone. Thus, after successful acquisition, the DPLLpreferably switches to any one of the data carrying bins and operates ina decision directed tracking mode. For relatively clean downstreamchannels, this approach is robust and bandwidth conserving.

From the foregoing description of the clock synchronization, it is seenthat all RSUs synchronize their receive clocks to the master clockprovided by the HE. The transmit clock used to synthesize thetransmissions from the transmitter of each RSU 30 is preferably likewisegenerated from the voltage controlled oscillator 240 at the RSU 30.Similarly, the receive clocks of the receiver 287 (see FIG. 7) of the HE25 may be generated from voltage controlled oscillator 190. The XMT VCXO190 itself is derived, for example, via an analog-PLL signal that, forexample, is received from a HE backplane. Thus, the HE 25 need onlyprovide a stable clock reference and the burden of synchronizationlargely rests with the RSUs 30.

This situation, however, is altered by any frequency shifts incurred inthe upstream transmissions from the RSUs 30 to the HE 25. Specifically,in some multipoint communication systems, frequency offsets areintroduced due to multiplexing of upstream signals from several remotesites onto a single optical fiber at intermediate nodes, for example,electro-optical nodes. The signals are demultiplexed at the HE 25 anddownconverted to the original sub-split return frequency band. Use ofsuch conversions can greatly reduce the noise level on the upstreamchannel along the transmission medium 35. However, the clocks used formultiplexing and de-multiplexing at these nodes are typically notsynchronized with each other (i.e. they are relatively free running) andintroduce frequency offsets, possibly in the range of several KHz. Sucha system is illustrated in FIG. 7.

The present inventors have recognized this problem and have provided asolution. With reference to FIG. 7, a second voltage controlledoscillator 290 is used at the HE 25 which serves as the clock referencefor one of the frequency synthesizers 295 included in the RF sub-systemof the receiver 287 of the HE 25. The voltage controlled oscillator 290has its output signal at line 300 synchronized with a pilot toneprovided by the transmitter 277 of the RSU 30. The pilot tone from theRSU 30 is provided in a predetermined bin. The HE 25 and RSU 30 needonly carry out the synchronization process once when the first RSUpowers up and is seeking to establish upstream communication with the HE25 or after bi-directional communications have been disrupted for aprolonged interval of time. Thereafter, the HE 25 maintains synchronismwith all RSU transmitters 277 by operating a symbol rate acquisitionDPLL in decision directed tracking mode. The use of this separatelysynchronized voltage controlled oscillator 290 compensates for frequencyoffsets introduced by the free running clocks of the multiplexers 310and demultiplexer 315. Note that in-spite of the upstream frequencyoffsets, the analog-to-digital converter 320 of the receiver 287 of theHE 25 and frequency synthesizer 325 used by mixer 30 can still be drivenoff the voltage controlled oscillator 190.

The upstream synchronization process is initiated after the RSU 30 hasacquired downstream synchronization and thereby has the capability toproperly receive messages from the HE 25. Concurrent with thetransmission of pilot tones, the HE demodulates all bins and continuallymeasures the received power in a predetermined window of a predeterminednumber of bins (possibly all bins designated for communication) around apre-designated bin. This bin, designated as the upstream pilot tone bin,is used by the first RSU 30 to transmit an upstream pilot tone afterderiving its own upstream clock via the synchronization processdescribed above. Since there is only one transmitting RSU 30, the HE 25can locate the pilot tone bin with relative ease, compute the frequencyoffset (if any), and appropriately adjust voltage controlled oscillator290.

One embodiment of the upstream synchronization process is set forth inconnection with FIG. 8. As illustrated, the synchronization processbegins with a search for the pilot tone transmitted by the RSU 30.Pursuant to the synchronization process, an initialization is firstperformed. In the initialization process, the transmitter 277 of the RSU30 transmits a pilot tone having constant phase and constant amplitudeafter it has successfully executed the downstream pilot tone search andacquisition processes. The pilot tone is transmitted in the upstreampilot tone bin for a predetermined number of consecutive symbols unlessit receives a message from the HE 25 to do otherwise.

If upstream synchronization is to proceed after initialization, the HE25 demodulates all bins and continuously measures receive power in awindow of a predetermined number of bins centered around thepre-designated upstream pilot tone bin. The HE 25 then locates the binfor which (a) receive power is maximum and (b) the magnitude ofdifference in power with adjacent bins exceeds a predeterminedthreshold. Such determinations can be made in the manner describedabove. Additionally, further coherency tests, etc., may also optionallybe employed. This bin is selected as the bin which is most likely tocontain the pilot tone. The HE 25 then uses the selected bin to computethe frequency offset (in bins) of the selected peak power bin from itsexpected location. Note that the offset would be zero when themuxing/demuxing process is ideal. The voltage controlled oscillator 290is then adjusted to compensate for the frequency offset by translatingthe offset frequency into a signed control voltage that is applied tothe voltage controlled oscillator 290 by a digital-to-analog converter330 (and, preferably, through a low pass filter 335) that is responsiveto a control signal provided by the digital signal processor 340. The HE25 then waits for a predetermined number of symbols and further refinesany adjustment that may be necessary until a zero offset results.

After the upstream pilot tone search process has been successfullycompleted, the HE 25 executes an upstream acquisition process. Thisacquisition process is substantially similar to the process noted abovein connection with the second pilot tone acquisition mode of the RSUreceiver 150. The DPLL parameters for the HE 25, however, are likely tobe different than those used in the RSU DPLL. After acquisition iscomplete, the HE 25 may send an appropriate message to the first RSU toindicate successful capture of and synchronization with the upstreampilot tone. At this point, the HE 25 and RSU 30 may be considered tohave successfully carried out sample timing and carrier recovery forboth the downstream and upstream channels.

After successful acquisition has taken place, the HE 25 needs toconstantly run the acquisition DPLL in a tracking mode. Loss of trackingcan lead to disruption in upstream communication with the entire RSUpopulation (often several hundred RSUs) served by the HE 25. Unlikeacquisition, tracking can reliably proceed in decision directed mode.The HE 25 instructs the first RSU to transmit random data in a bin lyingwithin the transmit range of the RSU 30. Even though the HE 25 employsrandom data transmissions from any one RSU 30, it can designate morethan one RSUs for additional reliability. This process continuesunchanged even if the RSUs being used for tracking begin to carry livetraffic.

Other aspects of an OFDM/DMT communications system are set forth in U.S.Ser. No. 08/845,544, filed Apr. 24, 1997, titled “SYMBOL ALIGNMENT IN AMULTIPOINT OFDM/DMT DIGITAL COMMUNICATION SYSTEM” and in co-pendingapplication Ser. No. 08/900,791, titled “APPARATUS AND METHOD ORUPSTREAM CLOCK SYNCHRONIZATION IN A MULTIPOINT OFDM/DMT DIGITALCOMMUNICATION SYSTEM”, that are hereby incorporated by reference. Theseother aspects, however, are not particularly pertinent to the presentsynchronization system.

Although the present invention has been described with reference tospecific embodiments, those of skill in the art will recognize thatchanges may be made thereto without departing from the scope and spiritof the invention as set forth in the appended claims.

1. An apparatus for synchronizing communications in an OFDM/TDMcommunications system, the apparatus comprising: a recovery module torecover a bin, including a pilot tone sub-symbol from a frequency rangeof a predetermined plurality of frequency bins, as a function of atleast one timing signal; an adjustment module to adjust the timingsignal as a function of phase differences measured between successivepilot tone sub-symbols; and a synchronization module to synchronizecommunications using the timing signal as adjusted to receive the pilottone sub-symbols within a frequency range sufficient for phase lockedloop processing of the pilot tone.
 2. The apparatus of claim 1 furtherincluding a generator module to generate OFDM/TDM symbols using at leastone transmitter timing signal, the at least one timing signal being usedfor timing inverse Fourier transform processing and carrier generationin transmitting the OFDM/DMT symbols, and a transmitter module totransmit the OFDM/TDM symbols over a predetermined number of bins acrossa transmission medium.
 3. The apparatus of claim 1 further including areceiver module to receive OFDM/DMT symbols from a transmission medium,the receiver including a demodulator to demodulate the OFDM/DMT symbolsusing the at least one timing signal, the at least one timing signalbeing used for timing Fourier transform processing and carriergeneration in demodulating the OFDM/DMT symbols.
 4. The apparatus ofclaim 1 wherein the recovery module is configured to compute coherencyof the pilot tone sub-symbols in each bin and reject bins havingsub-symbols with coherency below a predetermined threshold as notincluding the pilot tone sub-symbols.
 5. The apparatus of claim 1wherein the recovery module is configured to compute excessive coherencyof sub-symbols received in each bin having a coherency above apredetermined threshold and reject bins having sub-symbols havingexcessive coherency as not including the pilot tone sub-symbols.
 6. Theapparatus of claim 1 wherein the bin including the pilot tone sub-symbolis disposed adjacent to a plurality of proximate bins and wherein thebin containing the pilot tone sub-symbol and the plurality of proximatebins define a predetermined spectral pattern.
 7. The apparatus of claim1 wherein the recovery module is configured to identify a known spectralpattern and determine frequency of the bin including the pilot tonesub-symbol as a function of the spectral pattern to recover the binincluding the pilot tone sub-symbol.
 8. The apparatus of claim 7 whereinthe spectral pattern is symmetric.
 9. The apparatus of claim 7 whereinthe spectral pattern is asymmetric.
 10. The apparatus of claim 1 whereinthe synchronization module is arranged to adjust the timing signal as afunction of reducing differences between pilot sub-symbols over areceived series of frequency bins to synchronize the communication. 11.A method for synchronizing communication in an OFDM/TDM communicationssystem, the method comprising: recovering a bin including a pilot tonesub-symbol from a frequency range of a predetermined plurality offrequency bins as a function of a timing signal; and synchronizingcommunications as a function of phase differences measured betweensuccessive pilot tone sub-symbols; and adjusting the timing signal basedon the phase differences to receive the pilot tone sub-symbols within afrequency range sufficient for phase locked loop processing of the pilottone.
 12. The method of claim 11 further including generating OFDM/TDMsymbols using at least one transmitter timing signal, the timing signalbeing used for timing inverse Fourier transform processing and carriergeneration in transmitting the OFDM/DMT symbols and transmitting theOFDM/TDM symbols over a predetermined number of bins across atransmission medium.
 13. The method of claim 11 including receivingOFDM/DMT symbols from a transmission medium and demodulating theOFDM/DMT symbols using the at least one timing signal, the timing signalbeing used for timing Fourier transform processing and carriergeneration in demodulating received OFDM/DMT symbols.
 14. The method ofclaim 11 further including computing coherency of the pilot tonesub-symbols in each bin and rejecting bins having sub-symbols withcoherency below a predetermined threshold as not including the pilottone sub-symbols.
 15. The method of claim 11 further including computingexcessive coherency of sub-symbols received in each bin having acoherency above a predetermined threshold and rejecting bins havingsub-symbols having excessive coherency as not including the pilot tonesub-symbols.
 16. The method of claim 11 wherein the bin including thepilot tone sub-symbol is disposed adjacent to a plurality of proximatebins and the bin containing the pilot tone sub-symbol and the pluralityof proximate bins define a predetermined spectral pattern.
 17. Themethod of claim 11 further including identifying a known spectralpattern and determining frequency of the bin including the pilot tonesub-symbol as a function of the spectral pattern to recover the binincluding the pilot tone sub-symbol.
 18. The method of claim 17 whereinthe spectral pattern is symmetric.
 19. The method of claim 17 whereinthe spectral pattern is asymmetric.
 20. The method of claim 17 furtherincluding adjusting the timing signal as a function of reducingdifferences between pilot sub-symbols over a received series offrequency bins to synchronize the communication.